The present invention relates to a demodulator including an adaptive equalizer and a demodulating method, in digital communications.
In land mobile communications, out-of-sight radio communications are generally used. Accordingly, a received wave has a complex characteristic of being constituted by multiple waves which are much subjected to reflection, diffraction and scattering. In addition, the communication characteristic of a path channel varies instantaneously in accordance with the movement of a mobile station, so that communication quality is deteriorated. It is known that the variation in the communication characteristic generally accords with a Rayleigh distribution. This phenomenon is called "Rayleigh fading".
As a measure to cope with Rayleigh fading, particularly to cope with deterioration of communication quality caused by the instantaneous variation in amplitude, "diversity" is generally used. "Diversity" is a technique in which a received signal highest in received signal power among the several received signals is selected or synthesized from statistically independent received signals via a plurality of path channels to reduce the probability of lowering received signal power to thereby suppress the influence of Rayleigh fading. "Diversity" is classified into space diversity, directionality diversity, polarization diversity, etc., depending on the methods of selecting the independent received signal highest in received signal power. Further, "diversity" is classified into selection diversity, co-phasing combining diversity, maximal ratio combining diversity, etc., in accordance with the method of synthesizing independent received signals.
Because path channels constituting the above-mentioned plurality of path channels (multipath channels) have path channel lengths different from each other, the time of signal arrival at a reception point varies. The degree of this variation is called delay spread. The diversity is effective in the case where the delay spread is sufficiently small compared with transmission interval time per symbol, whereas bit error, so-called unrecoverable error, which is impossible to be compensated by only the diversity, occurs in the case where the delay spread is large.
FIG. 1 shows an example in which multipath channels in a base band are expressed by a model of time-variant filter. The reference numeral 1 designates a transmitted signal input terminal; 2-1, 2-2, . . . , 2-M, delay elements; 3-0, 3-1, 3-2, . . . , 3-M, coefficient multipliers; 4, an adder; and 5, a received signal output terminal. In FIG. 1, a transmitted symbol sequence supplied to the transmitted signal input terminal 1 is sent to the delay element 2-1 and the coefficient multiplier 3-0. Here, M delay elements 2-1, 2-2, . . . , 2-M (M is an integer) are series-connected and each has a delay time Ts equal to the transmission interval time of one symbol. The signals of the transmitted symbol sequence supplied to the transmitted signal input terminal 1 are sent to the delay elements 2-1, 2-2, . . . , 2-M successively. Signals are taken out from intermediate points between the delay elements 2-1 and 2-2, between the delay elements 2-2 and 2-3, . . . , between the delay elements 2-(M-1) and 2-M and from the last stage of the delay element 2-M, respectively, and supplied to the coefficient multipliers 3-1, 3-2, . . . , 3-M, respectively. The coefficient multipliers 3-0, 3-1, 3-2, . . . , 3-M have complex coefficients hi (i=0, 1, 2, . . . , M) which are set individually, so that the input signals are multiplied by the complex coefficients hi (i=0, 1, 2, . . . , M), respectively. Results of the multiplications are added up by the adder 4, so that the resulting value is sent to the received signal output terminal 5 and is output therefrom. As represented by the model in FIG. 1, a received signal at a certain point of time is provided as a result of addition of a direct wave component based on transmitted symbols at a corresponding point of time and a delay wave component constituted by transmitted symbols before the corresponding point of time.
When the delay spread is large, the received signal is subjected to intersymbol interference by the delay wave component in accordance with the case where any one of the complex coefficients except the complex coefficient h0 has a value of amplitude near the amplitude of the complex coefficient h0 or a plurality of complex coefficients except the complex coefficient h0 have values of amplitude near the amplitude of the complex coefficient h0.
The intersymbol interference due to the delay spread in such a manner is a cause of bit error which brings serious deterioration of communication quality. To suppress this deterioration, an adaptive equalizer, for example, represented by a decision feedback equalizer or a Viterbi equalizer is required to be used.
Adaptive equalizers are classified into linear equalizers and maximum likelihood sequence estimators (MLSE). The decision feedback equalizer is known as a representative example of the linear equalizers and the Viterbi equalizer is known as a representative example of the maximum likelihood sequence estimators.
Incidentally, a conventional example of the decision feedback equalizer has been described, for example, in J. G. Proakis, "Digital Communications", McGraw-Hill International Editions, 1989, pp. 593-600.
The decision feedback equalizer will be described below as an example of the adaptive equalizers. FIG. 2 is a diagram showing an example of the configuration of the decision feedback equalizer. The reference numerals 6 and 8 designate multipliers; 7, an adder; 11, a digital received signal input terminal; 12 and 13, delay elements; 15, a symbol decision unit; 16, a reference signal memory; 17, a switch; 18, an error estimator; 19, a tap coefficient update unit; 20, an equalization output terminal; 21, a feed-forward portion (FF portion); 22, a feedback portion (FB portion); and 23, an equalization filter. In FIG. 2, each of the delay elements 13 has a delay time Ts equal to the transmission interval time per symbol, and each of the delay elements 12 has a delay time Tp equal to a value which is given by the delay time of the delay element 13 divided by an integer (generally, by 2). Further, the equalization filter 23 is constituted by an FF portion 21, an FB portion 22, and an adder 7. The FF portion 21 is constituted by delay elements 12 and multipliers 6. The FB portion 22 is constituted by delay elements 13 and multipliers 8. Complex coefficients F-j, F-j+1, . . . , F0 called tap coefficients are set in the multipliers 6 respectively. Tap coefficients B1, B2, . . . , BK are set in the multipliers 8 respectively.
Received signals sampled at a sampling interval Tp are supplied to the digital received signal input terminal 11, sent to the delay elements 12 of the FF portion 21 of the equalization filter 23 and successively multiplied by corresponding tap coefficients through the multipliers 6 respectively. At the same time, output signals of the switch 17 are sent to the FB portion 22 and successively multiplied by corresponding tap coefficients through the multipliers 8 respectively. All the results of these multiplications are sent to the adder 7 and added up by the adder 7, so that the resulting signal of addition is provided as an output of the equalization filter 23. Not only the output signal of the equalization filter 23 is sent to the decision unit 15 and the error estimator 18 but also the output signal is taken out from the equalization output terminal 20. The decision unit 15 decides what symbol is expressed by the input signal and sends the decided symbol to one input of the switch 17.
In digital communications, a fixed symbol sequence is generally inserted for the purpose of synchronization, or the like. This symbol sequence, which is known also to the receiving side, is stored, as a reference signal, in the reference signal memory 16.
Synchronization symbols are generally used as the training symbol sequence stored in the reference signal memory 16. FIG. 3 shows an example of the structure of frame data used in digital communications. In digital communications, besides a symbol sequence corresponding to an information bit sequence to be transmitted, generally, synchronization words having fixed symbol patterns are inserted mainly for the purpose of establishing various kinds of synchronization, so that fixed-length data called "frame" which is formed of a combination of the synchronization words and the information symbol sequence is used as a unit to be transmitted. In this description, the switch 17 is operated at a point of time when the synchronization word of a head frame is sent, that is, by the start of the head frame.
Generally, the switch 17 selects the output of the symbol decision unit 15 and selects the output of the reference signal memory 16 only in a period in which the transmitted symbol is known. The output of the switch 17 is sent to the FB portion 22 and the error estimator 18. The error estimator 18 estimates error of the output of the equalization filter 23 with the output signal of the switch 17 as a reference and sends the estimated error to the tap coefficient update unit 19. The tap coefficient update unit 19 updates all tap coefficients of the equalization filter 23 individually at all times so that the estimated error supplied to the tap coefficient update unit 19 is converged into zero. As a result of the aforementioned operation, the influence of intersymbol interference, caused by the delay spread, on the signal taken out from the equalization output terminal 20 is reduced. Accordingly, the tap coefficients are updated so that the equalization filter 23 has a characteristic reverse to the characteristic of the time-variant filter which is a model of path channels described above with reference to FIG. 1. The tap coefficient update unit 19 uses adaptation algorithm for updating coefficients. Typical examples of the adaptation algorithm include steepest-decent gradient algorithm, least mean square (LMS) algorithm, recursive least square (RLS) algorithm, etc.
FIG. 4 is a graph for explaining an example of the operation of the decision unit 15 and the error estimator 18. In FIG. 4, the abscissa axis shows in-phase component I and the ordinate axis shows quadrature component Q. A quadrature phase shift keying (QPSK) modulation system will be described below with reference to FIG. 4. Dots 1, 2, 3 and 4 on the I-Q plane represent symbols. X points marked with a, b, . . . , h represent coordinates, on the I-Q plane, of input signals a, b, . . . , h supplied to the decision unit 15. Assuming now that an input signal a is supplied to the decision unit 15, the decision unit 15 detects the coordinate, on the I-Q plane, of the input signal a as the X point marked with a. Because the detected coordinate is in the first quadrant of the I-Q plane, the decision unit 15 outputs a symbol 1. Similarly, in the case where the input is b, c or d, the decision unit 15 outputs the symbol 1. In the case where the input is e or f, the decision unit 15 outputs a symbol 2. In the case where the input is g, the decision unit 15 outputs a symbol 3. In the case where the input is h, the decision unit 15 outputs a symbol 4. Then, the error estimator 18 calculates the difference between the symbol supplied from the decision unit 15 and the signal supplied from the adder 7 and supplies the difference to the tap coefficient update unit 19. For example, when the output supplied from the adder 7 is a, the error estimator 18 outputs Ea in FIG. 4. Similarly, when the output supplied from the adder 7 is b, c, . . . , h, the error estimator 18 outputs Eb, Ec, . . . , Eh in FIG. 4.
As described above, a receiver strong against the instantaneous variation caused by Rayleigh fading and delay spread of a path channel characteristic can be formed by combination of "diversity" and an adaptive equalizer. The same rule as described above applies to other linear equalizers and maximum likelihood sequence estimators such as Viterbi equalizers.
As described above, the adaptive equalizer used in mobile radio communications requires a tracking characteristic in accordance with the instantaneous variation in the path channel characteristic. On the other hand, it is desired to avoid the influence of noise contained in the received signal. These tracking characteristic and noise resistance are characteristics about updating of tap coefficients but, generally, they have a trade-off relation with each other. That is, if a tracking characteristic is enhanced so as to be adapted to the higher-speed variation in the path channel characteristic caused by high-speed movement or high carrier frequency, the influence of noise becomes so large that error occurs in tap coefficients to bring adaptive equalizing error. Accordingly, in addition to noise contained in the received signal and intersymbol interference caused by delay spread, deterioration of equalization caused by this adaptive equalizing error is also required to be taken into consideration as a cause of communication deterioration. When balance between the quantity of improvement of intersymbol interference based on equalization and the quantity of deterioration of equalization under a noisy environment is taken into consideration, to use an adaptive equalization output in spite of unnecessity originally in the case where delay spread is so small that the influence of intersymbol interference is small, is to bring a cause of communication deterioration instead. Therefore, it is necessary to control the adaptive equalizing operation in accordance with the size of the delay spread. It may be said that this problem was apt to be overlooked conventionally because attention was paid to a point that a receiver was configured so as to be adapted to the case where the geographical condition of a mobile station was assumed to be most severe.